Transmission circuit and method for transmitting a bit sequence to be transmitted

ABSTRACT

A transmission circuit and method for transmitting a bit sequence to be transmitted is provided that includes a dividing device for dividing the bit sequence into an even-numbered bit sequence and into an odd-numbered bit sequence, a first device for forming the first sampled values of a first fundamental wave depending on the even-numbered bit sequence, a second device for forming second sampled values of a second fundamental wave depending on the odd-numbered bit sequence, the second fundamental wave being shifted relative to the first fundamental wave by a time period, and includes a summator for summation of the first sampled values of the first fundamental wave and the second sampled values of the second fundamental wave to form an output value sequence.

This nonprovisional application claims priority to German Patent Application No. 10 2008 027 389.9, which was filed in Germany on Jun. 9, 2008, and to U.S. Provisional Application No. 61/060,619, which was filed on Jun. 11, 2008, and which are both herein incorporated by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a transmission circuit and a method for transmitting a bit sequence to be transmitted.

2. Description of the Background Art

A binary phase transformation (BPSK: binary phase shift keying) for the 868/915 MHz band with a data rate of 20 kb/s in the 868 MHz band and 40 kb/s in the 915 MHz Band is known from the industry standard IEEE 802.15.4. Spreading using two spreading codes (binary) and pulse shaping occurs in a transmission method. Bits in a bit sequence here have precisely one position, so that several bits are not transmitted in parallel.

SUMMARY OF THE INVENTION

It is therefore an object of the present invention to improve as much as possible a method for transmitting a bit sequence. Instead of explicit binary spreading with precisely two code words and pulse shaping, an alternative method for transmitting a bit sequence to be transmitted is provided.

The bit sequence can be divided into an even-numbered bit sequence and an odd-numbered bit sequence. For this purpose, an even-numbered bit and an odd-numbered bit are assigned alternately accordingly to the even-numbered bit sequence and the odd-numbered bit sequence.

Depending on the even-numbered bit sequence, first sampled values of a first fundamental wave are formed. Depending on the odd-numbered bit sequence, second sampled values of a second fundamental wave are formed. The first fundamental wave and the second fundamental wave are thereby defined by the totality of the respective sampled values in regard to the specific bit.

The second fundamental wave is shifted by a time period relative to the first fundamental wave. The time period of the shift can be equal to the bit time of a bit. The first sampled values and the second sampled values can be equal in each case for a value {−1 or 1} of the bits.

The first sampled values of the first fundamental wave and the second sampled values of the second fundamental wave can be added to form an output value sequence.

The output value sequence is then processed. Advantageously, the rate of sampling of the output value sequence is converted. Preferably, the digital values of the output value sequence are converted digital-to-analog and then filtered. Advantageously, a carrier signal modulation, dependent on the output value sequence, occurs for the transmission.

The invention has as its object, furthermore, to provide as improved a transmission circuit as possible. Accordingly, a transmission circuit for transmitting a bit sequence to be transmitted is provided.

The transmission circuit can have a dividing device for dividing the bit sequence into an even-numbered bit sequence and into an odd-numbered bit sequence.

The transmission circuit has a first device for forming first sampled values of a first fundamental wave depending on the even-numbered bit sequence.

The transmission circuit has a second device for forming second sampled values of a second fundamental wave depending on the odd-numbered bit sequence. The second fundamental wave is shifted by a time period relative to the first fundamental wave.

The transmission circuit can have a summator for summation of the first sampled values of the first fundamental wave and the second sampled values of the second fundamental wave, so that the output value sequence is formed as a sum.

The invention furthermore has as its object to provide a use. Accordingly, a use of sampled values readable from a memory are provided to form an output value sequence, which represents in approximation spreading and pulse shaping of a bit sequence. The output value sequence is converted digital-to-analog within the scope of a binary chip sequence for transmitting the bit sequence.

The refinements described hereinafter relate to the transmission circuit, as well as to the use and to the method for transmission.

According to an embodiment, to form the first sampled values, the first sampled values are read out depending on a bit value of the even-numbered bit sequence. Preferably, the first sampled values are read out of a memory. To form the second sampled values the second sampled values are read out depending on a bit value of the odd-numbered bit sequence. The second sampled values are read out of a memory. For reading-out, addresses, for example, can be output to a memory. Advantageously, sampled values can be output by multiplexers for reading-out. It is also possible to control or clock a state machine for reading out the sampled values.

It is provided in another embodiment that the first sampled values are read out via a control of a first multiplexer, said control being dependent on the bit value of the even-numbered bit sequence. Alternatively or in combination, the second sampled values are read out by means of a control of a second multiplexer, said control being dependent on the bit value of the odd-numbered bit sequence.

For shifting the second fundamental wave relative to the first fundamental wave by the time period, the second multiplexer can be controlled offset by a number of sampled values relative to the first multiplexer.

Alternatively, for shifting the second fundamental wave relative to the first fundamental wave by the time period, it is advantageous to delay the second sampled values by means of a delay element.

According to an embodiment, it is provided that the sampling rate can be increased after the summation of the first sampled values and the second sampled values by sample rate conversion.

According to an embodiment, the first device can have a first multiplexer. The first multiplexer can be connected to a memory for reading out the first sampled values. The second device can have a second multiplexer. The second multiplexer can be connected to the memory for reading out the second sampled values. The memory is, for example, a read-only memory, whose stored values are formed, for example, by hard wiring with logic zero or with logic one (ROM). Alternatively, the memory is programmable and has nonvolatile memory cells, for example, (EEPROM).

According to an embodiment, it is provided that the first multiplexer and/or the second multiplexer can be formed for shifting the second fundamental wave relative to the first fundamental wave. For example, the control of the multiplexer is offset by the number of samplings forming the shift.

In an alternative embodiment, a delay element can be connected downstream of the second device for shifting the second fundamental wave relative to the first fundamental wave.

The dividing device can be formed as a (for example, third) multiplexer. The dividing device is preferably formed to assign the bit values (−1, 1) of the bit sequence alternately to the even-numbered bit sequence and the odd-numbered bit sequence by multiplexing.

According to an embodiment, a sample rate converter can be connected downstream of the summator to increase the sampling rate.

In another embodiment, it is provided that the first sampled values and the second sampled values correspond to spreading and pulse shaping. The sampled values can be calculated according to the spreading and pulse shaping.

Further scope of applicability of the present invention will become apparent from the detailed description given hereinafter. However, it should be understood that the detailed description and specific examples, while indicating preferred embodiments of the invention, are given by way of illustration only, since various changes and modifications within the spirit and scope of the invention will become apparent to those skilled in the art from this detailed description.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention will become more fully understood from the detailed description given hereinbelow and the accompanying drawings which are given by way of illustration only, and thus, are not limitive of the present invention, and wherein:

FIG. 1 is a first schematic block diagram;

FIG. 2 is a second schematic block diagram according to an embodiment;

FIG. 3 is a third schematic block diagram according to an embodiment; and

FIG. 4 is a fourth schematic block diagram according to an embodiment.

DETAILED DESCRIPTION

Pulse amplitude modulation in conjunction with frequency spreading methods, such as DSSS (direct sequence spread spectrum), can be used in communication technology systems. The signal bandwidth is increased considerably with the aid of spreading codes compared with the information rate. Advantages of such spreading, inter alia, are increased robustness to narrow-band interference and multipath propagation.

In FIG. 1, the information-bearing bit sequence b(kT_(b)) is limited to the alphabetic values of {−1, +1} (binary) and therefore not multidigit. Here, T_(b) is the bit time. Each bit is now assigned precisely one corresponding code word from the binary spreading code {c⁻¹, c₊₁}, with

c ⁻¹=(c ⁻¹ ⁰ ,c ⁻¹ ¹ , . . . , c ⁻¹ ^(N−1))  (1)

c ₊₁=(c ₊₁ ⁰ ,c ₊₁ ¹ , . . . , c ₊₁ ^(N−1))  (2)

The coefficients c^(i) ⁻¹⁺¹ of the code words (chips) in this case are any, also optionally complex-valued, signal constellations. The associated chip sequence c^(i) _(b(kTb)) is filtered at the output of encoder 110 with a pulse filter 120 with pulse response g(t) relative to the chip time T_(c)=T_(b)/N. The baseband signal in the case of pulse-amplitude modulation (PAM) arises as:

$\begin{matrix} {{y(t)} = {\sum\limits_{k\;}^{\;}{\sum\limits_{j = 0}^{N - 1}{c_{b{({kT}_{b})}}^{j}{g\left( {t - {jT}_{c} - {kT}_{b}} \right)}}}}} & (3) \end{matrix}$

In FIG. 1, an implementation form of DSSS with time-discrete filtering is shown schematically as a block diagram. The chip sequence c^(i) _(b(kTb)) is converted by sample rate conversion (SRC) to a higher rate 1/T_(z), whereby the pulse shaping may be part of the sample rate conversion. The sequence y(mT_(z)) forms approximately the baseband signal y(t) by means of digital-to-analog converter 130 (DAC) in conjunction with a continuous-time anti-aliasing filter 140 (AAF). With a sufficiently high rate 1/T_(z), the technical complexity of anti-aliasing filter 140 (AAF) declines.

The following FIGS. 2 through 4 show simplified implementations of pulse amplitude modulation (PAM) in conjunction with binary spreading codes.

The simplified implementation is based on the basic idea that the baseband signal can be divided into components in regard to an even bit sequence and an odd bit sequence:

$\begin{matrix} {{y(t)} = {{\sum\limits_{k\;}^{\;}{\sum\limits_{j = 0}^{N - 1}{c_{b{({2\; {kT}_{b}})}}^{j}{g\left( {t - {jT}_{c} - {2\; {kT}_{b}}} \right)}}}} + {\sum\limits_{k\;}^{\;}{\sum\limits_{j = 0}^{N - 1}{c_{b{({{({{2\; k} + 1})}T_{b}})}}^{j}{g\left( {t - {jT}_{c} - {\left( {{2\; k} + 1} \right)T_{b}}} \right)}}}}}} & (4) \end{matrix}$

Because the fundamental frequency of the pulse shaping filter is specified in regard to 1/T_(c), the following applies in good approximation for code lengths that are not too small N>>2:

$\begin{matrix} {{{{g(t)}} \approx {0\mspace{14mu} {for}\mspace{14mu} {t}} \geq \frac{T_{b}}{2}} = {\frac{N}{2}T_{c}}} & (5) \end{matrix}$

Thus, the following applies:

$\begin{matrix} {{y(t)} \approx {{\sum\limits_{j = 0}^{N - 1}{c_{b{({2\; {kT}_{b}})}}^{j}{g\left( {t - {jT}_{c} - {2\; {kT}_{b}}} \right)}}} + {\sum\limits_{j = 0}^{N - 1}{c_{b{({{({{2\; k} + 1})}T_{b}})}}^{j}{g\left( {t - {jT}_{c} - {\left( {{2\; k} + 1} \right)T_{b}}} \right)}}}}} & (6) \end{matrix}$

The baseband signal at time t therefore forms approximately by a superimposition of two fundamental waves shifted in time by the bit time T_(b)

$\begin{matrix} {{y_{w}^{b}(t)} = {\sum\limits_{j = 0}^{N - 1}{c_{b}^{j}{{g\left( {t - {jT}_{c}} \right)}.}}}} & (7) \end{matrix}$

where each fundamental wave can only assume the functions of {y_(w) ⁻¹(t),y_(w) ⁺¹(t)}. By elimination of the summation over k, the fundamental waves of the odd bit sequence and the even bit sequence are memoryless, so that no temporary storage of values is necessary. In this case, the fundamental waves are finite because of condition (5), i.e., limited to 2T_(b).

Instead of a device that provides for explicit spreading with subsequent sample rate conversion, a schematic block diagram of a transmission circuit with DSSS and pulse shaping by superimposition of fundamental waves offset in time is shown in FIG. 2.

The bit sequence b(kT_(b)) is divided into an even-numbered bit sequence b(2kT_(b)), therefore the bits with the index 0, 2, 4, 6, 8, etc., and into an odd-numbered bit sequence b((2k+1)T_(b)), therefore the bits with the index 1, 3, 5, 7, etc. A multiplexer 230 is provided for this purpose, which assigns the bit values of the bit sequence b(kT_(b)) alternately to the even-numbered bit sequence b(2kT_(b)) and the odd-numbered bit sequence b((2k+1)T_(b)) by switching.

Sampled values y^(b) _(w)(nT_(z)−T_(b)/2) are output by a first device 210 and a second device 220. For the output, devices 210, 220 have, for example, a state machine or multiplexer for selecting, for example, hard-wired values. The first device 210 in this case is formed to output sampled values as a function of the even-numbered bit sequence b(2kT_(b)) and as a function of a counter n. The second device 220 in this case is formed to output sampled values as a function of the odd-numbered bit sequence b((2k+1)T_(b)) and as a function of a counter n. A control unit 250 is provided for this purpose.

The sampled values output by second device 220 are delayed by means of a delay element 270 with the function z^(−M/2) by the time duration T_(b), therefore the time duration of a bit. The output sampled values y1 of first device 210 and sampled values y2, output by second device 220 and delayed by delay element 270, are added by summator 260. The sum forms the output value sequence y(mT_(z)). A digital-to-analog converter 280, which converts the output value sequence y(mT_(z)) into analog values, is connected downstream of summator 260. The analog values are filtered by filter 290 (AAF: anti-aliasing filter), connected downstream of digital-to-analog converter 280, to obtain the baseband signal y(t).

Instead of a circuit that provides explicit spreading with subsequent sample rate conversion, in the exemplary embodiment of FIG. 2, the sampled values of the digital sampled values y1, y2, offset in time by T_(b), of the fundamental values, offset in time, are output and then added to form the digital values y(mT_(z)). The sampled values y1, y2 of the fundamental waves for b=−1 and b=+1 are stored in each case preferably in a table.

Because of condition (5), the length of the fundamental wave is limited approximately to 2T_(b). For a fixed b, the

M=2T _(b) /T _(z)  (8)

values y^(b) _(w)(nT_(z)−T_(b)/2) for

n=(m modM)=0, . . . , M−1  (9)

are needed. In this case, m goes from minus infinite to plus infinite. n determines the current sampled value of the fundamental wave.

The time shifting by T_(b), therefore by M/2 sampled values, naturally need not be realized absolutely in the form of delay elements, such as 270 in FIG. 2. In the exemplary embodiment of FIG. 3, instead the relative delay of the fundamental waves is generated by the control of a first multiplexer 310 (MUX₀) and a second multiplexer 320 (MUX₁) by means of control device 350. First multiplexer 310 and second multiplexer 320 are connected to a memory unit, for example, a hard-wired value logic (ROM) or alternatively to a programmable memory unit (EEPROM) with the values y^(b) _(w)(nT_(z)−T_(b)/2) stored, for example, in tables. The generation here occurs again as a function of an even-numbered bit sequence b(2kT_(b)) and an odd-numbered bit sequence b((2k+1)T_(b)) generated by a third multiplexer 330.

The two multiplexers 310 and 320 in this case are operated independently of one another. First multiplexer 310 forms the values y^(b) _(w)(nT_(z)−T_(b)/2) as a function of n and b=b(2kT_(b)). The values output by second multiplexer 320 are shifted cyclically by M/2 values relative to first multiplexer 310. The switching of second multiplexer 320 occurs relative to b=b((2k+1)T_(b)). Again, the sampled values y1, y2 are obtained, which are added to the output sequence y(mT_(z)) by summator 360. The output sequence y(mT_(z)) is converted by digital-to-analog-converter 380 into an analog signal, which is filtered by means of filter 390 and output as baseband signal y(t).

FIG. 4 shows another exemplary embodiment. Multiplexers 410, 420, 430, a control circuit 450, a summator 460, a digital-to-analog-converter 480, and a filter 490 are again provided. In this exemplary embodiment, instead of the target rate R_(z) first an intermediate rate

$\begin{matrix} {R_{z}^{\prime} = {\frac{1}{l}R_{z}}} & (10) \end{matrix}$

with an integer />1 is set to generate values y′(m′T_(z)′). The number of inputs into the multiplexers and number of table values are reduced as a result by the factor /. The remaining increase in rate to the required rate R_(z) at the input of digital-to-analog converter 480 occurs by an additional sampled rate increase by means of a sample rate converter 465. Compared with the transmission circuit of FIG. 1, the total outlay (for a not too large /) is much lower, because the interpolation filter at the transition R_(z)−>R_(z) can be substantially more broadband compared with R_(c)−>R_(z). In this regard, the effect occurs that recursive filter structures can be used for an efficient implementation. Another effect is that because of the integrity of /, it is to be assumed that clocking of the rate R_(z)′ can be derived in simple form from the rate R_(z).

In fact, the previously described embodiment variants can be used in principle with any system with binary spreading. Preferably, the previously described embodiment variants, however, are used with a system according to the industry standard IEEE 802.15.4. In this case, binary phase shift keying is provided. The binary phase shift keying (BPSK) occurs in the 868/915 MHz bands. The data rate in the 868 MHz band is 20 kb/s and in the 915 MHz band 40 kb/s. In this case, a spreading procedure DSSS with BPSK for modulation of the chip is carried out. A differential coding is provided for the data symbol encoding. Each bit from the PPDU (physical layer protocol data unit) is processed by differential coding, bit-to-chip mapping, and modulation functions in an octet-like sequence, beginning with the preamble field and ending with the last octet of the PSDU (physical layer service data unit). For each octet, the bit with the lowest significance is processed first and the bit with the highest significance last.

The differential coding is connected upstream of multiplexers 230, 330, or 430 for use of the industry standard IEEE 802.15.4, but not shown in FIGS. 2 to 4. This is a modulo-2 addition, therefore an exclusive-or function of the current data bit and the previously differentially coded bit.

A 15-chip PN sequence is provided for the binary bit-to-chip mapping:

Input bit chip values ^(c0 . . .) ^(c14) −1 1 1 1 1 −1 1 −1 1 1 −1 −1 1 −1 −1 −1 1 −1 −1 −1 −1 1 −1 1 −1 −1 1 1 −1 1 1 1

The chip sequences are modulated on a carrier signal by means of BPSK. In this case, raised cosine pulse shaping with the roll-off factor of 1 is used. In this case, a chip value of 1 is assigned to a positive pulse and a chip value of −1 to a negative pulse. The chip with the lowest value is transmitted first and the chip with the highest value last.

For the binary bit-to-chip mapping and BPSK modulation, the exemplary embodiments of FIGS. 2 to 4 enable significant simplification of the circuit without negatively affecting the functionality of the system defined by the industry standard IEEE 802.15.4.

The invention is not limited to the shown embodiment variants of FIGS. 2 through 4 and also not to the industry standard IEEE 802.15.4. For example, it is also possible to use the embodiment variants for a different transmission standard with binary spreading. It is also possible to divide the even-numbered bit sequence and/or the odd-numbered bit sequence into additional bit sequences, whereby accordingly additional fundamental waves shifted to one another are formed.

The invention being thus described, it will be obvious that the same may be varied in many ways. Such variations are not to be regarded as a departure from the spirit and scope of the invention, and all such modifications as would be obvious to one skilled in the art are to be included within the scope of the following claims. 

1. A method for transmitting a bit sequence to be transmitted, the method comprising: dividing the bit sequence into an even-numbered bit sequence and into an odd-numbered bit sequence; forming first sampled values of a first fundamental wave based on the even-numbered bit sequence; forming second sampled values of a second fundamental wave based on the odd-numbered bit sequence, the second fundamental wave being shifted by a time period relative to the first fundamental wave; and adding the first sampled values of the first fundamental wave and the second sampled values of the second fundamental wave to form an output value sequence.
 2. The method according to claim 1, wherein, to form the first sampled values, the first sampled values are read out based on a bit value of the even-numbered bit sequence, and wherein, to form the second sampled values, the second sampled values are read out depending on a bit value of the odd-numbered bit sequence.
 3. The method according to claim 2, wherein the first sampled values are read out via a control of a first multiplexer, the control being dependent on the bit value of the even-numbered bit sequence and/or the second sampled values are read out via a control of a second multiplexer, the control being dependent on the bit value of the odd-numbered bit sequence.
 4. The method according to claim 1, wherein, for shifting the second fundamental wave relative to the first fundamental wave by the time period, the second multiplexer is controlled offset by a number of sampled values relative to the first multiplexer.
 5. The method according to claim 1, wherein for shifting the second fundamental wave relative to the first fundamental wave by the time period, the second sampled values are delayed by a delay element.
 6. The method according to claim 1, wherein the sampling rate is increased after the summation of the first sampled values and the second sampled values.
 7. A transmission circuit for transmitting a bit sequence to be transmitted, the transmission circuit comprising: a dividing device configured to divide the bit sequence into an even-numbered bit sequence and into an odd-numbered bit sequence; a first device configured to form the first sampled values of a first fundamental wave depending on the even-numbered bit sequence; a second device configured to form second sampled values of a second fundamental wave based on the odd-numbered bit sequence, the second fundamental wave being shifted relative to the first fundamental wave by a time period; and a summator configured to sum the first sampled values of the first fundamental wave and the second sampled values of the second fundamental wave to form an output value sequence.
 8. The transmission circuit according to claim 7, wherein the first device has a first multiplexer, the first multiplexer is connectable to a memory for reading out the first sampled values, the second device has a second multiplexer, and the second multiplexer is connectable to the memory for reading out the second sampled values.
 9. The transmission circuit according to claim 7, wherein the first multiplexer and/or the second multiplexer are formed for shifting the second fundamental wave relative to the first fundamental wave.
 10. The transmission circuit according to claim 7, wherein a delay element is connected downstream of the second device for shifting the second fundamental wave relative to the first fundamental wave.
 11. The transmission circuit according to claim 7, wherein the dividing device is a third multiplexer assigned the bit values of the bit sequence alternately to the even-numbered bit sequence and the odd-numbered bit sequence by multiplexing.
 12. The transmission circuit according to claim 7, wherein a sample rate converter is connected downstream of the summator to increase the sampling rate.
 13. The transmission circuit according to claim 7, wherein the first sampled values and the second sampled values correspond to spreading and pulse shaping.
 14. Use of sampled values readable from a memory to form an output value sequence, the output value sequence representing, in approximation, spreading and pulse shaping of a binary bit sequence, wherein the output value sequence within the scope of a binary chip sequence is converted digital-to-analog for transmitting the bit sequence. 